Circuit technique for high speed low power data transfer bus

ABSTRACT

A high speed low power data transfer bus circuit that reduces bus power consumption by imposing a limited, controlled voltage swing on the associated data bus. In one embodiment, an inverter is coupled with a pMOS pass transistor and an nMOS discharge transistor, and the combination is coupled with a data bus. The discharge transistor and pass transistor can be programmed to provide a preselected bus operational characteristics. In another embodiment, multiple nMOS discharge transistors can be coupled to the data bus via the pass transistor, with each of the discharge transistors being selectively programmed to provide additional preselected bus operational characteristics, multiple, programmable discharge transistors, thus selectably imposing encoded and multilevel logic signals on the data bus. In another embodiment, a bidirectional data transfer bus circuit couples two data busses while imposing a limited, controlled voltage swing during the transfer. One preferred embodiment couples a first data bus and a second data bus with cross-linked inverters. Interposed between the inverters, and its associated bus, is a respective pMOS pass transistor. Also, coupled between each input node and ground, is a signal discharge transistor, preferably nMOS, which facilitates data transfer between the buses. Each of the inverters is coupled with a clocked charge/discharge circuit, preferably using a common clock signal.

CROSS-REFERENCE TO RELATED APPLICATION(S)

The present application claims the benefit of the filing dates of thefollowing United States Provisional Patent Applications, the contents ofall of which are hereby expressly incorporated herein by reference:

Ser. No. 60/215,741, filed Jun. 29, 2000, and entitled MEMORY MODULEWITH HIERARCHICAL FUNCTIONALITY;

Ser. No. 60/193,607, filed Mar. 31, 2000, and entitled MEMORY REDUNDANCYIMPLEMENTATION;

Ser. No. 60/193,606, filed Mar. 31, 2000, and entitled DIFFUSION REPLICADELAY CIRCUIT;

Ser. No. 60/179,777, filed Feb. 2, 2000, and entitled SPLIT DUMMYBITLINES FOR FAST, LOW POWER MEMORY;

Ser. No. 60/193,605, filed Mar. 31, 2000, and entitled A CIRCUITTECHNIQUE FOR HIGH SPEED LOW POWER DATA TRANSFER BUS;

Ser. No. 60/179,766, filed Feb. 2, 2000, and entitled FAST DECODER WITHASYNCHRONOUS RESET;

Ser. No. 60/220,567, filed Jul. 25, 2000, and entitled FAST DECODER WITHROW REDUNDANCY;

Ser. No. 60/179,866, filed Feb. 2, 2000, and entitled HIGH PRECISIONDELAY MEASUREMENT CIRCUIT;

Ser. No. 60/179,718, filed Feb. 2, 2000, and entitled LIMITED SWINGDRIVER CIRCUIT;

Ser. No. 60/179,765, filed Feb. 2, 2000, and entitled SINGLE-ENDED SENSEAMPLIFIER WITH SAMPLE-AND-HOLD REFERENCE;

Ser. No. 60/179,768, filed Feb. 2, 2000, and entitled SENSE AMPLIFIERWITH OFFSET CANCELLATION AND CHARGE-SHARE LIMITED SWING DRIVERS; and

Ser. No. 60/179,865, filed Feb. 2, 2000, and entitled MEMORYARCHITECTURE WITH SINGLE PORT CELL AND DUAL PORT (READ AND WRITE)FUNCTIONALITY.

The following related patent applications, assigned to the same assigneehereof and filed on even date herewith in the names of the sameinventors as the present application, disclose related subject matter,with the subject of each being incorporated by reference herein in itsentirety:

Memory Module with Hierarchical Functionality, Ser. No. 09/775,477; HighPrecision Delay Measurement Circuit, Ser. No. 09/776,262; Single-EndedSense Amplifier with Sample-and-Hold Reference, Ser. No. 09/776,220;Limited Switch Driver Circuit, Ser. No. 09/775,478; Fast Decoder withAsynchronous Reset with Row Redundancy; Ser. No. 09/775,476; DiffusionReplica Delay Circuit, Ser. No. 09/776,029; Sense Amplifier with OffsetCancellation and Charge-Share Limited Swing Drivers, Ser. No.09/775,475; Memory Architecture with Single-Port Cell and Dual-Port(Read and Write) Functionality, Ser. No. 09/775,701 and MemoryRedundancy Implementation, Ser. No. 09/776,263.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to memory devices, in particular,semiconductor memory devices, and most particularly, scalable,power-efficient semiconductor memory devices.

2. Background of the Art

Memory structures have become integral parts of modern VLSI systems,including digital signal processing systems. Although it typically isdesirable to incorporate as many memory cells as possible into a givenarea, memory cell density is usually constrained by other design factorssuch as layout efficiency, performance, power requirements, and noisesensitivity.

In view of the trends toward compact, high-performance, high-bandwidthintegrated computer networks, portable computing, and mobilecommunications, the aforementioned constraints can impose severelimitations upon memory structure designs, which traditional memorysystem and subcomponent implementations may fail to obviate.

One type of basic storage element is the static random access memory(SRAM), which can retain its memory state without the need forrefreshing as long as power is applied to the cell. In an SRAM device,the memory state II usually stored as a voltage differential within abistable functional element, such as an inverter loop. A SRAM cell ismore complex than a counterpart dynamic RAM (DRAM) cell, requiring agreater number of constituent elements, preferably transistors.Accordingly, SRAM devices commonly consume more power and dissipate moreheat than a DRAM of comparable memory density, thus efficient;lower-power SRAM device designs are particularly suitable for VLSIsystems having need for high-density SRAM components, providing thosememory components observe the often strict overall design constraints ofthe particular VLSI system. Furthermore, the SRAM subsystems of manyVLSI systems frequently are integrated relative to particular designimplementations, with specific adaptions of the SRAM subsystem limiting,or even precluding, the scalability of the SRAM subsystem design. As aresult SRAM memory subsystem designs, even those considered to be“scalable”, often fail to meet design limitations once these memorysubsystem designs are scaled-up for use in a VLSI system with need for agreater memory cell population and/or density.

There is a need for an efficient, scalable, high-performance, low-powermemory structure that allows a system designer to create a SRAM memorysubsystem that satisfies strict constraints for device area, power,performance, noise sensitivity, and the like. Moreover, there is a needfor a high speed low power data transfer bus.

SUMMARY OF THE INVENTION

The present invention satisfies the above needs by providing a highspeed low power data transfer bus circuit that reduces bus powerconsumption by imposing a limited, controlled voltage swing on theassociated data bus. In one embodiment of the present invention, thecontrolled voltage swing data bus circuit, an inverter is coupled with apass transistor, preferably pMOS, and a discharge transistor, preferablynMOS, and the combination is coupled with a data bus. The dischargetransistor is preferably programmed to provide a first preselected busoperational characteristic by controlling the rate and extent of voltagedecay on the data bus, which can occur when the pass transistor is ON,coupling the discharge transistor through to the data bus. The passtransistor also may be programmed to enhance the control of a busoperational characteristic. In another embodiment of the inventionherein, multiple discharge transistors, again, preferably nMOS, can becoupled to the data bus via the pass transistor. In this case, it ispreferred that each of the discharge transistors be selectivelyprogrammed to provide additional preselected bus operationalcharacteristics by selectably controlling the rate and extent of voltagedecay on the data bus. Advantageously, by employing multiple,programmable discharge transistors, multiple, distinct logic levels canbe selectably imposed on the data bus. In addition, the availability ofmultilevel logic permits transferring encoded data to the data bus.Concurrently with effecting a reduction in power consumption, limitedvoltage swings on the data bus tends to increase the speed of the bus.Furthermore, multilevel logic signal can transmit information with fewersignal lines, relative to standard bi-level logic signals.

In another embodiment of the invention herein, a bidirectional highspeed low power data transfer bus circuit couples two data busses whileimposing a limited, controlled voltage swing during the transfer. Again,power consumption is reduced even while increasing the speed of the bus.One preferred embodiment of the bidirectional data transfer bus couplesa first data bus and a second data bus with cross-linked inverters.Interposed between each of the inverters, and its associated bus, is arespective pass transistor, preferably pMOS, with the node between thepass transistor and the inverter forming the input node for therespective bus. Also, coupled between each input node and ground, is asignal discharge transistor, preferably nMOS, which facilitates datatransfer between the buses.

Furthermore, it is desirable to couple each of the inverters with aclocked charge/discharge circuit, preferably using a common clocksignal, which charge/discharge circuit can precharge/discharge the inputnodes, depending upon the state of the clock and the data on theassociated bus. While it is preferred to program the signal dischargetransistors to provide preselected bus operational characteristics,including for example, rate of voltage decay on the associated bus,additional programmed signal discharge transistors may be included inthe bidirectional circuit to effect multilevel logic.

The data transfer bus circuits of the present invention can be used tocouple a sense amplifier or a wordline decoder, including for example, aglobal sense amplifier, a local sense amplifier, a global wordlinedecoder, and a local wordline decoder, to a data bus.

The present invention will be more fully understood from the followingdetailed description of the embodiments thereof, taken together with thefollowing drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features, aspects and advantages of the presentinvention will be more fully understood when considered with respect tothe following detailed description, appended claims and accompanyingdrawings, wherein:

FIG. 1 is a block diagram of an exemplary static random access memory(SRAM) architecture;

FIG. 2 is a general circuit schematic of an exemplary six-transistorCMOS SRAM memory cell;

FIG. 3 is a block diagram of an embodiment of a hierarchical memorymodule using local bitline sensing, according to the present invention;

FIG. 4 is a block diagram of an embodiment of a hierarchical memorymodule using an alternative local bitline sensing structure;

FIG. 5 is a block diagram of an exemplary two-dimensional, two-tierhierarchical memory structure, employing plural local bitline sensingmodules of FIG. 3;

FIG. 6 is a block diagram of an exemplary hierarchical memory structuredepicting a memory module employing both local word line decoding andlocal bitline sensing structures;

FIG. 7 is a perspective illustration of a hierarchical memory structurehaving a three-tier hierarchy, in accordance with the invention herein;

FIG. 8 is a circuit schematic of an asynchronously-resettable decoder,according to an aspect of the present invention;

FIG. 9 is a circuit schematic of a limited swing driver circuit,according to an aspect of the present invention;

FIG. 10 is a circuit schematic of a single-ended sense amplifier circuitwith sample-and-hold reference, according to an aspect of the presentinvention;

FIG. 11 is a circuit schematic of charge-share, limited-swing driversense amplifier circuit, according to an aspect of the presentinvention;

FIG. 12 is a block diagram illustrating an embodiment of hierarchicalmemory module redundancy;

FIG. 13 is a block diagram illustrating another embodiment ofhierarchical memory module redundancy;

FIG. 14 is a block diagram of a memory redundancy device, illustratingyet another embodiment of hierarchical memory module redundancy;

FIG. 15A is a diagrammatic representation of the signal flow of anexemplary unfaulted memory module featuring column-oriented redundancy;

FIG. 15B is a diagrammatic representation of the shifted signal flow ofthe exemplary faulted memory module illustrated in FIG. 15A;

FIG. 16 is a generalized block diagram of a redundancy selector circuit,illustrating still another embodiment of hierarchical memory moduleredundancy;

FIG. 17 is a circuit schematic of an embodiment of a global row decoderhaving row redundancy according to the invention herein;

FIG. 18 is a block diagram illustrating dual-port functionality in asingle-port hierarchical memory structure employing hierarchical memorymodules according to the present invention;

FIG. 19 is a schematic diagram of one embodiment of a high precisiondelay measurement circuit, according to the present invention;

FIG. 20 is a simplified block diagram of one aspect of the presentinvention employing one embodiment of a diffusion replica delay circuit;

FIG. 21 is a simplified block diagram of one aspect of the presentinvention employing another embodiment of a diffusion replica delaycircuit;

FIG. 22A is a schematic diagram of another aspect of an embodiment ofthe present invention, employing a high-speed, low-power data transferbus circuit; and

FIG. 22B is a schematic diagram of another aspect of an embodiment ofthe present invention, employing a high-speed, low-power data transferbus circuit.

DETAILED DESCRIPTION OF THE EMBODIMENTS

As will be understood by one having skill in the art, most VLSI systems,including communications systems and DSP devices contain VLSI memorysubsystems. Modern applications of VLSI memory subsystems almostinvariably demand high efficiency, high performance implementations thatmagnify the design tradeoff between layout efficient, speed, powerconsumption, scalability, design tolerances, and the like. The presentinvention ameliorates these tradeoffs using a novel hierarchicalarchitecture. The memory module of the present invention also can employone or more novel components which further add to the memory modulesefficiency and robustness.

Hereafter, but solely for the purposes of exposition, it will be usefulto describe the various aspects and embodiments of the invention hereinin the context of an SRAM memory structure, using CMOS SRAM memorycells. However, it will be appreciated by those skilled in the art thepresent invention is not limited to CMOS-based processes and that,mutatis mutandi, these aspects and embodiments may be used in categoriesof memory products other than SRAM, including without limitation, DRAM,ROM, PLA, and the like, whether embedded within a VLSI system, or astand alone memory device.

Exemplary SRAM Module and Storage Cell

FIG. 1 is a functional block diagram of SRAM memory structure 100 thatillustrates the basic features of most SRAM subsystems. Module 100includes memory core 102, word line controller 104, precharge controller112, memory address inputs 114, and bitline controller 116. Memory core102 is composed of a two-dimensional array of K-bits of memory cells103, which is arranged to have C columns and R rows of bit storagelocations, where K=[C×R] . The most common configuration of memory core102 uses single word line 106 to connect cells 103 onto paireddifferential bitlines 118. In general, core 102 is arranged as an arrayof 2^(P) word lines, based on a set of P memory address input lines 114i.e., R=2^(P). Thus, the p-bit address is decoded by row address decoder110 and column address decoder 122. Access to a given memory cell 103within such a single-core memory is accomplished by activating thecolumn 105 and the row 106 corresponding to cell 103. Column 105 isactivated by selecting, and switching, all bitlines in the particularcolumn corresponding to cell 103.

The particular row to be accessed is chosen by selective activation ofrow address decoder 110, which usually corresponds uniquely with a givenrow, or word line, spanning all cells 103 on the particular row. Also,word driver 108 can drive selected word line 106 such that selectedmemory cell 103 can be written into or read out, on a particular pair ofbitlines 118, according to the bit address supplied to memory addressinputs 114.

Bitline controller 116 can include precharge cells 120, columnmultiplexers 122, sense amplifiers 124, and input/output buffers 126.Because differential read/write schemes are typically used for memorycells, it is desirable that bitlines be placed in a well-defined statebefore being accessed. Precharge cells 120 can be used to set up thestate of bitlines 118, through a PRECHARGE cycle, according to apredefined precharging scheme. In a static precharging scheme, prechargecells 120 can be left continuously on. While often simple to implement,static precharging can add a substantial power burden to active deviceoperation. Dynamic precharging schemes can use clocked precharge cells120 to charge the bitlines and, thus, can reduce the power budget ofstructure 100. In addition to establishing a defined state on bitlines118, precharging cells 120 can also be used to effect equalization ofdifferential voltages on bitlines 118 prior to a read operation. Senseamplifiers 124 allow the size of memory cell 103 to be reduced bysensing the differential voltage on bitline 118, which is indicative ofits state, and translating that differential voltage into a logic-leversignal.

In general a READ operation is performed by enabling row decoder 110,which selects a particular row. The charge on one bitlines 118 from eachpair of bitlines on each column will discharge through the enabledmemory cell 103, representing the state of the active cells 103 on thatcolumn 105. Column decoder 122 will enable only one of the columns, andwill connect bitlines 118 to input/output buffer 126. Sense amplifiers124 provide the driving capability to source current to input/outputbuffer 126. When sense amplifier 124 is enabled, the unbalanced bitlines118 will cause the balanced sense amplifier to trip toward the state ofthe bitlines, and data 125 will be output by buffer 126.

A WRITE operation is performed by applying data 125 to I/O buffers 126.Prior to the WRITE operation, bitlines 118 are precharged by prechargecells 120 to a predetermined value. The application of input data 125 toI/O buffers 126 tend to discharge the precharge voltage on one of thebitlines 118, leaving one bitline logic HIGH and one bitline logic LOW.Column decoder 122 selects a particular column 105 connecting bitlines118 to I/O buffers 126, thereby discharging one of the bitlines 118. Therow decoder 110 selects a particular row, and the information onbitlines 118 will be written on cell 103 at the intersection of column105 and row 106. At the beginning of a typical internal timing cycle,precharging is disabled, and is not enabled again until the entireoperation is completed. Column decoder 122 and row decoder 110 are thenactivated, followed by the activation of sense amplifier 124. At theconclusion of a READ or a WRITE operation, sense amplifier 124 isdeactivated. This is followed by disabling decoders 110, 122, at whichtime precharge cells 120 become active again during a subsequentPRECHARGE cycle. In general, keeping sense amplifier 124 activatedduring the entire READ/WRITE operation leads to excessive device powerconsumption, because sense amplifier 124 needs to be active only for theactual time required to sense the state of memory cell 103.

FIG. 2 illustrates one implementation of memory cell 103 in FIG. 1, inthe form of six-transistor CMOS cell 200. Transistor cell 200 is onetype of transistor which also may be used in embodiments of the presentinvention. SRAM cell 200 can be in one of three possible states: (1) theSTABLE state, in which cell 200 holds a signal value corresponding to alogic “1” or logic “0”; (2) a READ operation state; or (3) a WRITEoperation state. In the STABLE state, memory cell 200 is effectivelydisconnected from the memory core (e.g., core 102 in FIG. 1). Bitlines202, 204 are precharged HIGH (logic “1”) before any operation (READ orWRITE) can take place. Row select transistors 206, 208 are turned offduring precharge. Precharge power is supplied by precharge cells (notshown) coupled with the bitlines 202, 204, similar to precharge cells120 in FIG. 1. A READ operation is initiated by performing a PRECHARGEcycle, precharging bitlines 202, 204 to logic HIGH, and activating wordline 205 using row select transistors 206, 208. One of the bitlines 202,204 discharges through bit cell 200, and a differential voltage is setupbetween the bitlines 202, 204. This voltage is sensed and amplified tologic levels. A WRITE operation to cell 200 is carried out after anotherPRECHARGE cycle, by driving bitlines 202, 204 to the required state, andactivating word line 205. CMOS is a desirable technology because thesupply current drawn by such an SRAM cell typically is limited to theleakage current of transistors 201 a-d while in the STABLE state.

As memory cell density increases, and as memory components are furtherintegrated into more complex systems, it becomes imperative to providememory architectures that are robust, reliable, fast, and area- andpower-efficient. Single-core architectures, similar to those illustratedin FIG. 1, are increasingly unable to satisfy the power, speed, area androbustness constraints for a given high-performance memory application.Therefore, it is desirable to minimize power consumption, increasedevice speed, and improve device reliability and robustness, andnumerous approaches have been developed to those ends. The advantages ofthe present invention may be better appreciated within the followingcontext of some of these approaches, particularly as they relate topower reduction and speed improvement, and to redundancy and robustness.

Power Reduction and Speed Improvement

In reference to FIG. 1, the content of memory cell 103 of memory block100 is detected in sense amplifier 102, using a differential signalbetween bitlines 104, 106. However, this architecture is not scalable.Also, as memory block 100 is made larger, there are practicallimitations to the ability of sense amplifier 102 to receive an adequatesignal in a timely fashion at bitlines 104, 106. Increasing the lengthof bitlines 104, 106, increases the associated bitline capacitance and,thus, increases the time needed for a signal to develop on bitlines 104,106. More power must be supplied to lines 104, 106 to overcome theadditional capacitance. Also, under the architectures of the existingart, it takes more time to precharge longer bitlines, thereby reducingthe effective device speed. Similarly, writing to longer bitlines 104,106, as found in the existing art, requires more extensive precharging,thereby increasing the power demands of the circuit, and furtherreducing the effective device speed.

In general, reduced power consumption in memory devices such asstructure 100 in FIG. 1 can be accomplished by, for example, reducingtotal switched capacitance, and minimizing voltage swings. Theadvantages of the power reduction aspects of certain embodiments of thepresent invention can further be appreciated within the context ofswitched capacitance reduction and voltage swing limitation.

Switched Capacitance Reduction

As the bit density of memory structures increases, it has been observedthat single-core memory structures can have unacceptably large switchingcapacitances associated with each memory access. Access to any bitlocation within such a single-core memory necessitates enabling theentire row, or word line, in which the datum is stored, and switchingall bitlines in the structure. Therefore, it is desirable to designhigh-performance memory structures to reduce the total switchedcapacitance during any given access.

Two well-known approaches for reducing total switched capacitance duringa memory structure access include dividing a single-core memorystructure into a banked memory structure, and employing divided wordline structures. In the former approach, it is necessary to activateonly the particular memory bank associated with the memory cell ofinterest. In the latter approach, total switched capacitance is reducedby localizing word line activation to the greatest practicable extent.

Divided or Banked Memory Core

One approach to reducing switching capacitances is to divide the memorycore into separately switchable banks of memory cells. Typically, thetotal switched capacitance during a given memory access for bankedmemory cores is inversely proportional to the number of banks employed.By judiciously selecting the number and placement of bank units within agiven memory core design, as well as the type of decoding used, thetotal switching capacitance, and thus the overall power consumed by thememory core, can be greatly reduced. A banked design also may realize ahigher product yield, because the memory banks can be arranged such thata defective bank is rendered inoperable and inaccessible, while theremaining operational banks of the memory core can be packed into alower-capacity product.

However, banked designs may not be appropriate for certain applications.Divided memory cores demand additional decoding circuitry to permitselective access to individual banks, and incur a delay as a result.Also, many banked designs employ memory segments that are merelyscaled-down versions of traditional monolithic core memory designs, witheach segment having dedicated control, precharging, decoding, sensing,and driving circuitry. These circuits tend to consume much more power inboth standby and operational modes, than do their associated memorycells. Such banked structures may be simple to design, but theadditional complexity and power consumption thus can reduce overallmemory component performance.

By their very nature, banked designs are not suitable for scaling-up toaccommodate large design requirements. Also, traditional banked designsmay not be readily conformable to applications requiring a memory coreconfiguration that is substantially different from the underlying memorybank architecture (e.g., a memory structure needing relatively few rowsof very long bit-length word lengths). Rather than resort to a top-downdivision of the basic memory structure using banked memory designs,preferred embodiments of the present invention provide a hierarchicalmemory structure that is synthesized using a bottom-up approach, byhierarchically coupling basic memory modules with localizeddecision-making features that synergistically cooperate to dramaticallyreduce the overall power needs, and improve the operating speed, of thestructure. At a minimum, such a basic hierarchical module can includelocalized bitline sensing.

Divided Word Line

Often, the bit-width of a memory component is sized to accommodate aparticular word length. As the word length for a particular designincreases, so do the associated word line delays, switched capacitance,power consumption, and the like. To accommodate very long word lines, itmay be desirable to divide core-spanning global word lines into localword lines, each consisting of smaller groups of adjacent, word-orientedmemory cells. Each local group employs local decoding and drivingcomponents to produce the local word line signals when the global wordline, to which it is coupled, is activated. In long word lengthapplications, the additional overhead incurred by divided word lines canbe offset by reduced word line delays, power consumption and so forth.However, the added overhead imposed by existing divided word lineschemes may make it unsuitable for many implementations. As before,rather than resorting to the traditional top-down division of wordlines, certain preferred embodiment of the invention herein includeproviding a local word line to the aforementioned basic memory module,which further enhances the local decision making features of the module.As before, by using a bottom-up approach to hierarchically couple basicmemory modules, here with the added localized decision-making featuresof local word lines according to the present invention, additionalsynergies are realized, which further reduce overall power consumptionand signal propagation times.

Voltage-swing Reduction Techniques

Power reduction also can be achieved by reducing the voltage swingsexperienced throughout the structure. By limiting voltage swings, it ispossible to reduce the amount of power dissipated as the voltage at anode or on a line decays during a particular event or operation, as wellas to reduce the amount of power required to return the various decayedvoltages to the desired state after the particular event or operation,or prior to the next access. Two techniques to this end include usingpulsed word lines and sense amplifier voltage swing reduction.

Pulsed Word Lines

By enabling a word line just long enough to correctly detect thedifferential voltage across a selected memory cell, it is possible toreduce the bitline voltage discharge corresponding to a READ operationon the selected cell. In some designs, by applying a pulsed signal tothe associated word line over a chosen interval, a sense amplifier isactivated only during that interval, thereby reducing the duration ofthe bitline voltage decay. These designs typically use some form ofpulse generator that produces a fixed-duration pulse. If the duration ofthe pulse is targeted to satisfy worst-case timing scenarios, theadditional margin will result in unnecessary bitline current draw duringnominal operations. Therefore, it is desirable to employ a self-timed,self-limiting word line device that is responsive to the actual durationof a given READ operation on a selected cell, and that substantiallylimits word line activation to that duration. Furthermore, where a senseamplifier can successfully complete a READ operation in less than amemory system clock cycle, it also may be desirable that the pulse widthactivation be asynchronous, relative to the memory system clock. Certainaspects of the present invention provide a pulsed word line signal, forexample, using a cooperative interaction between global and local wordline decoders.

Sense Amplifier Voltage Swing Reduction

In order to make large memory arrays, it is most desirable to keep thesize of an individual memory cell to a minimum. As a result, individualmemory cells generally are incapable of supplying driving current toassociated input/output bitlines. Sense amplifiers typically are used todetect the value of the datum stored in a particular memory cell and toprovide the current needed to drive the I/O lines. In sense amplifierdesign, there typically is a trade-off between power and speed, withfaster response times usually dictating greater power requirements.Faster sense amplifiers can also tend to be physically larger, relativeto low speed, low power devices. Furthermore, the analog nature of senseamplifiers can result in their consuming an appreciable fraction of thetotal power. Although one way to improve the responsiveness of a senseamplifier is to use a more sensitive sense amplifier, any gainedbenefits are offset by the concomitant circuit complexity whichnevertheless suffers from increased noise sensitivity. It is desirable,then, to limit bitline voltage swings and to reduce the power consumedby the sense amplifier.

In one typical design, the sense amplifier detects the smalldifferential signals across a memory cell, which are in an unbalancedstate representative of datum value stored in the cell, and amplifiesthe resulting signal to logic level. Prior to a READ operation, thebitlines associated with a particular memory column are precharged to achosen value. When a specific memory cell is enabled, a row decoderselects the particular row in which the memory cell is located, and anassociated column decoder selects a sense amplifier associated with theparticular column. The charge on one of those bitlines is dischargedthrough the enabled memory cell, in a manner corresponding to the valueof the datum stored in the memory cell. This produces an imbalancebetween the signals on the paired bitlines, and causing a bitlinevoltage swing. When enabled, the sense amplifier detects the unbalancedsignal and, in response, the usually-balanced sense amplifier statechanges to a state representative of the value of the datum. This statedetection and response occurs within a finite period, during which aspecific amount of power is dissipated. The longer it takes to detectthe unbalanced signal, the greater the voltage decay on the prechargedbitlines, and the more power dissipated during the READ operation. Anypower that is dissipated beyond the actual time necessary for sensingthe memory cell state, is truly wasted power. In traditional SRAMdesigns, the sense amplifiers that operate during a particular READoperation, remain active during nearly the entire read cycle. However,this approach unnecessarily dissipates substantial amounts of power,considering that a sense amplifier needs to be active just long enoughto correctly detect the differential voltage across a selected memorycell, indicating the stored memory state.

There are two general approaches to reducing power in sense amplifiers.First, sense amplifier current can be limited by using sense amplifiersthat automatically shut off once the sense operation has completed. Onesense amplifier design to this end is a self-latching sense amplifier,which turns off as soon as the sense amplifier indicates the senseddatum state. Second, sense amplifier currents can be limited byconstraining the activation of the sense amplifier to precisely theperiod required. This approach can be realized through the use of adummy column circuit, complete with bit cells, sense amplifier, andsupport circuitry. By mimicking the operation of a functional column,the dummy circuit can provide to a sense amplifier timing circuit anapproximation of the activation period characteristic of the functionalsense amplifiers in the memory system. Although the dummy circuitapproximation can be quite satisfactory, there is an underlyingassumption that all functional sense amplifiers have completed thesensing operation by the time the dummy circuit completes the itsoperation. In that regard, use of a dummy circuit can be similar toenabling the sense amplifiers with a fixed-duration pulsed signal.Aspects of the present invention provide circuitry and sense amplifierswhich limit voltage swings, and which improve the sensitivity androbustness of sense amplifier operation. For example, compact,power-conserving sense amplifiers having increased immunity to noise, aswell as to intrinsic and operational offsets, are provided. In thecontext of the present invention, such sense amplifiers can be realizedat the local module tier, as well as throughout the higher tiers of ahierarchical memory structure, according to the present invention.

Redundancy

Memory designers typically balance power and device area against speed.High-performance memory components place a severe strain on the powerand area budgets of associated systems particularly where suchcomponents are embedded within a VLSI system, such as a digital signalprocessing system. Therefore, it is highly desirable to provide memorysubsystems that are fast, yet power-and area-efficient. Highlyintegrated, high performance components require complex fabrication andmanufacturing processes. These processes experience unavoidableparameter variations which can impose physical defects upon the unitsbeing produced, or can exploit design vulnerabilities to the extent ofrendering the affected units unusable, or substandard.

In a memory structure, redundancy can be important, for example, becausea fabrication flaw, or operational failure, of even a single bit cellmay result in the failure of the system relying upon the memory.Likewise, process invariant features may be needed to insure that theinternal operations of the structure conform to precise timing andparametric specifications. Lacking redundancy and process invariantfeatures, the actual manufacturing yield for a particular memorystructure can be unacceptably low. Low-yield memory structures areparticularly unacceptable when embedded within more complex systems,which inherently have more fabrication and manufacturingvulnerabilities. A higher manufacturing yield translates into a lowerper-unit cost and robust design translates into reliable products havinglower operational costs. Thus, it is also highly desirable to designcomponents having redundancy and process invariant features whereverpossible.

Redundancy devices and techniques constitute other certain preferredaspects of the invention herein which, alone or together, enhance thefunctionality of the hierarchical memory structure. The aforementionedredundancy aspects of the present invention can render the hierarchicalmemory structure less susceptible to incapacitation by defects duringfabrication or during operation, advantageously providing a memoryproduct that is at once more manufacturable and cost-efficient, andoperationally more robust. Redundancy within a hierarchical memorymodule can be realized by adding one or more redundant rows, columns, orboth, to the basic module structure. In one aspect of the presentinvention a decoder enabling row redundancy is provided. Moreover, amemory structure composed of hierarchical memory modules can employ oneor more redundant modules for mapping to failed memory circuits. Aredundant module can provide a one-for-one replacement of a failedmodule, or it can provide one or more memory cell circuits to one ormore primary memory modules.

Memory Module with Hierarchical Functionality

The modular, hierarchical memory architecture according to the inventionherein provides a compact, robust, power-efficient, high-performancememory system having, advantageously, a flexible and extensivelyscalable architecture. The hierarchical memory structure is composed offundamental memory modules which can be cooperatively coupled, andarranged in multiple hierarchical tiers, to devise a composite memoryproduct having arbitrary column depth or row length. This bottom-upmodular approach localizes timing considerations, decision making, andpower consumption to the particular unit(s) in which the desired data isstored.

Within a defined design hierarchy, the fundamental memory modules can begrouped to form a larger memory block, that itself can be coupled withsimilar memory structures to form still larger memory blocks. In turn,these larger structures can be arranged to create a complex structure atthe highest tier of the hierarchy. In hierarchical sensing, it isdesired to provide two or more tiers of bit sensing, thereby decreasingthe read and write time of the device, i.e., increasing effective devicespeed, while reducing overall device power requirements. In ahierarchical design, switching and memory cell power consumption duringa read/write operation are localized to the immediate vicinity of thememory cells being evaluated or written, i.e., those memory cells inselected memory modules, with the exception of a limited number ofglobal word line selectors and sense amplifiers, and support circuitry.The majority of modules that do not contain the memory cells beingevaluated or written generally remain inactive.

Preferred embodiments of the present invention provide a hierarchicalmemory module using local bitline sensing, local word line decoding, orboth, which intrinsically reduces overall power consumption and signalpropagation, and increases overall speed, as well as design flexibilityand scalability. Aspects of the present invention contemplate apparatusand methods which further limit the overall power dissipation of thehierarchical memory structure, while minimizing the impact of amulti-tier hierarchy. Certain aspects of the present invention aredirected to mitigate functional vulnerabilities that may develop fromvariations in operational parameters, or that related to the fabricationprocess. In addition, devices and techniques are disclosed whichadvantageously ameliorate system performance degradation resulting fromtemporal inefficiencies, including, without limitation, a high-precisiondelay measurement circuit, a diffusion delay replication circuit andassociated dummy devices. In another aspect of the present invention, anasynchronously resettable decoder is provided that reduces the bitlinevoltage discharge, corresponding, for example, to a READ operation onthe selected cell, by limiting word-line activation to the actual timerequired for the sense amplifier to correctly detect the differentialvoltage across a selected memory cell.

Hierarchical Memory Modules

In prior art memory designs, such as the aforementioned banked designs,large logical memory blocks are divided into smaller, physical modules,each having the attendant overhead of an entire block of memoryincluding predecoders, sense amplifiers, multiplexers, and the like. Inthe aggregate, such memory blocks would behave as an individual memoryblock. However, using the present invention, memory blocks ofcomparable, or much larger, size can be provided by couplinghierarchical functional modules into larger physical memory blocks ofarbitrary number of words and word length. For example, existing designswhich aggregate smaller memory blocks into a single logical blockusually require the replication of the predecoders, sense amplifiers,and other overhead circuitry that would be associated with a singlememory block. According to the present invention, this replication isunnecessary, and undesirable. One embodiment of the inventioncomprehends local bitline sensing, in which a limited number of memorycells are coupled with a single local sense amplifier, thereby forming abasic memory module. Similar memory modules are grouped and arranged tooutput the local sense amplifier signal to the global sense amplifiersignal. Thus, the bitlines associated with the memory cells are notdirectly coupled with a global sense amplifier, mitigating the signalpropagation delay and power consumption typically associated with globalbitline sensing. In this approach, the local bitline sense amplifierquickly and economically sense the state of a selected memory cell andreport the state to the global sense amplifier. In another embodiment ofthe invention herein, the delays and power consumption of global wordline decoding are mitigated by providing a memory module, composed of alimited number of memory cells, having local word line decoding. Similarto the local bitline sensing approach, a single global word line decodercan be coupled with the respective local word line decoders of multiplemodules. When the global decoder is activated with an address, only thelocal word line decoder associated with the desired memory cellresponds, and activates the memory cell. This aspect, too, isparticularly power-conservative and fast, because the loading on theglobal line is limited to the associated local word line decoders, andthe global word line signal need be present only as long as required totrigger the relevant local word line. In yet another embodiment of thepresent invention, a hierarchical memory module employing both localbitline sensing and local word line decoding is provided, which realizesthe advantages of both approaches. Each of the above embodiments arediscussed forthwith.

Local Bitline Sensing

FIG. 3 illustrates a memory block 300 formed by coupling multiplecooperating constituent modules 320 a-e, with each of the modules 320a-e having a respective local sense amplifier 308 a-e. Each module iscomposed of a predefined number of memory cells 325 a-g, which arecoupled with one of the respective local sense amplifiers 308 a-e. Eachlocal sense amplifiers 308 a-e is coupled with global sense amplifier302 via bitlines 304, 306. Because each of local sense amplifiers 308a-e sense only the local bitlines 310 a-e, 312 a-e, of the respectivememory modules 320 a-e, the amount of time and power necessary toprecharge local bitlines 310 a-e and 312 a-e are substantially reduced.Only when local sense amplifier 308 a-e senses a signal on respectivelocal lines 310 a-e and 312 a-e, does it provide a signal to globalsense amplifier 302. This architecture adds flexibility and scalabilityto a memory architecture design because the memory size can be increasedby adding locally-sensed memory modules such as 320 a-e.

Increasing the number of local sense amplifiers 308 a-e attached toglobal bitlines 304, 306, does not significantly increase the loadingupon the global bitlines, or increase the power consumption in globalbitlines 304, 306 because signal development and precharging occur onlyin the local sense amplifier 308 a-e, proximate to the signal found inthe memory cells 325 a-g within corresponding memory module 320 a-e.

In preferred embodiments of the invention herein, it is desirable tohave each module be self-timed. That is, each memory module 320 a-e canhave internal circuitry that senses and establishes a sufficient periodfor local sensing to occur. Such self-timing circuitry is well-known inthe art. In single-core designs, or even banked designs, self-timingmemory cores may be unsuitable for high-performance operation, becausethe timing tends to be dependent upon the slowest of many components inthe structure, and because the signal propagation times in such largestructures can be significant. The implementation of self-timing inthese larger structures can be adversely affected by variations infabrication and manufacturing processes, which can substantially impactthe operational parameters of the memory array and the underlying timingcircuit components.

In a hierarchical memory module, self-timing is desirable because thetiming paths for each module 320 a-e comprehends only a limited numberof memory cells 325 a-g over a very limited signal path. Each module, ineffect, has substantial autonomy in deciding the amount of time requiredto execute a given PRECHARGE, READ, or WRITE operation. For the mostpart, the duration of an operation is very brief at the local tier,relative to the access time of the overall structure, so that memorystructure 300 composed of hierarchical memory modules 320 a-e is notsubject to the usual difficulties associated with self-timing, and alsois resistant to fabrication and manufacturing process variations.

In general, the cores of localized sense amplifiers 308 a-e can besmaller than a typical global sense amplifier 302, because a relativelylarger signal develops within a given period on the local senseamplifier bitlines, 310 a-e, 312 a-e. That is, there is more signalavailable to drive local sense amplifier 308 a-e. In aglobal-sense-amplifier-only architecture, a greater delay occurs while asignal is developed across the global bitlines, which delay can bedecreased at the expense of increased power consumption. Advantageously,local bit sensing implementations can reduce the delay whilesimultaneously reducing consumed power.

In certain aspects of the invention herein, detailed below, a limitedswing driver signal can be sent from the active local sense amplifier tothe global sense amplifier. A full swing signal also may be sent, inwhich case, a very simple digital buffer, may be used. However, if alimited swing signal is used, a more complicated sense amplifier may beneeded. For a power constrained application, it may be desirable toshare local sense amplifiers among two or more memory modules. Senseamplifier sharing, however, may slightly retard the bit signaldevelopment line indirectly because, during the first part of a sensingperiod, the capacitances of each of the top and the bottom shared memorymodules are being discharged. However, this speed decrease can beminimized and is relatively small, when compared to the benefits gainedby employing logical sense amplifiers over the existing global-onlyarchitectures. Moreover, preferred embodiments of the invention hereincan obviate these potentially adverse effects of sense amplifier sharingby substantially isolating the local sense amplifier from associatedlocal bitlines which are not coupled with the memory cell to be sensed.

FIG. 4 shows a memory structure 400, which is similar to structure 300in FIG. 3, by providing local bitline sensing of modules 420 a-d. Eachmemory module 420 a-d is composed of a predefined number of memory cells425 a-g. Memory cells 425 a-gare coupled with respective local senseamplifier 408 a, b via local bitlines 410 a-d, 412 a-d. Unlike structure300 in FIG. 3, where each module 320 a-e has its own local senseamplifier 308 a-e, memory modules 420 a-d are paired with a single senseamplifier 408 a, b. Similar to FIG. 3, FIG. 4 shows global senseamplifier 402 being coupled with local sense amplifiers 408 a, 408 b.

FIG. 5 further illustrates that memory structures such as module 300 inFIG. 3 can be coupled such that the overall structure is extended inaddress size (this is vertically), or in bit length (this ishorizontally), or both. The arrayed structure in FIG. 5 also can usemodules such as module 400 in FIG. 4. FIG. 5 also illustrates that acomposite memory structure 500 using hierarchical memory modules can betruly hierarchical. Memory blocks 502, 503 can be composed of multiplememory modules, such as module 504, which can be modules as described inreference to FIG. 3 and FIG. 4. Each memory block 502, 503 employstwo-tier sensing, as previously illustrated. However, in structure 500,memory blocks 502, 503 employ an intermediate tier of bitline sensing,using, for example, midtier sense amplifiers 514, 516. Under thehierarchical memory paradigm, midtier sense amplifiers 514, 516 can becoupled with global sense amplifier 520. Indeed, the hierarchical memoryparadigm, in accordance with the present invention, can comprehend ahighly-scalable multi-tiered hierarchy, enabling the memory designer todevise memory structures having memory cell densities and configurationsthat are tailored to the application. Advantageously, this scalabilityand configurability can be obtained without the attendant delays, andsubstantially increased power and area consumption of prior art memoryarchitectures.

One of the key factors in designing a faster, power-efficient device isthat the capacitance per unit length of the global bitline can be madeless than the capacitance of the local bitlines. This is because, usingthe hierarchical scheme, the capacitance of the global bitline is nolonger constrained by the cell design. For example, metal lines can berun on top of the memory device. Also, a multiplexing scheme can be usedthat increase the pitch of the bitlines, thereby dispersing them,further reducing bitline capacitance. Overall, the distance between theglobal bitlines can be wider, because the memory cells are not directlyconnected to the global bitlines. Instead, each cell, e.g. cell 303 inFIG. 3., is connected only to the local sense amplifier, e.g. senseamplifier 308 a-e.

Local Word Line Decoding

FIG. 6 illustrates a hierarchical structure 600 having hierarchicalword-line decoding in which each hierarchical memory module 605 iscomposed of a predefined number of memory cells 610, which are coupledwith a particular local word line decoder 615 a-c. Each local word linedecoder 615 a-c is coupled with a respective global word line decoder620. Each global word line decoder 620 a-d is activated when predecoder622 transmits address information relevant to a particular global wordline decoder 620 a-d via predecoder lines 623. In response, global wordline decoder 620 a-d activates global word line 630 which, in turn,activates a particular local word line decoder 615 a-c. Local word linedecoder 615 a-c then enables associated memory module 605, so that theparticular memory cell 610 of interest can be evaluated. Each of memorymodules 605 can be considered to be an independent memory component tothe extent that the hierarchical functionality of each of modules 605relies upon local sensing via local sense amplifiers 608 a-b, localdecoding via local word line decoders 615 a-c, or both. As with otherpreferred embodiments of the invention herein, it is desirable to haveeach module 605 be self-timed. Self-timing can be especially useful whenused in conjunction with local word line decoding because a local timingsignal from a respective one of memory module 605 can be used toterminate global word line activation, local bitline sensing, or both.

Similar to the scaling illustrated in FIG. 5, multiple memory devices600 can be arrayed coupled with global bitlines or global decoding wordlines, to create a composite memory component of a desired size andconfiguration. In an embodiment of the present invention, 256 rows ofmemory are used in each module 605, allowing the memory designer tocreate a memory block of arbitrary size, having a 256 row granularity.For prior art memory devices, a typical realistic limitation to thenumber of bits sense per sense amplifier is about 512 bit. Long bit orword lines can present a problem, particularly for a WRITE operations,because the associated driver can be limited by the amount of power itcan produce, and the speed at which sufficient charge can be built-upupon signal lines, such as global bitlines 604, 606 in FIG. 6.

Although FIG. 6 shows hierarchical word line decoding used inconjunction with hierarchical bitline operations, hierarchical word-linedecoding can be implemented without hierarchical bitline sensing. It ispreferred to use both the hierarchical word line decoding, and thehierarchical bitline sensing to obtain the synergistic effects ofdecreased power and increased speed for the entire device.

Hierarchical Functionality

In typical designs, power intends to increase approximately linearlywith the size of the memory. However, according to the presentinvention, as illustrated in FIG. 3 through FIG. 6, power requirementsmay increase only fractionally as the overall memory structure sizeincreases, primarily because only the memory module, and associatedlocal bitlines and local word lines are activated during a givenoperation. Due to the localized functionality, the global bitlines andword lines are activated for relatively brief periods at the beginningand end of the operation. In any event, power consumption is generallydictated by the bit size of the word, and the basic moduleconfiguration, i.e., the number of rows and row length of modules 620a-e. Thus, significant benefits can be realized by judiciously selectingthe configuration of a memory module, relative to the overall memorystructure configuration. For example, in a memory structure according tothe present invention, a doubling in the size of the memory device canaccount for power consumption increase of about twenty percent, and nota doubling, as found in prior art designs. Furthermore, a memorystructure according to the present invention can realize afour-to-six-fold decrease in power requirements and can operate 30% to50% faster, and often more, than traditional architectures.

FIG. 7 illustrates that memory structures according to the presentinvention, for example memory structure 740, are fully hierarchical, inthat each tier within the hierarchy includes local bit line sensing,local word line decoding, or both. Exemplary memory structure 740 isthree-tier hierarchical device with memory module 700 beingrepresentative of the fundamental, or lowest, tier (L₀) of the memoryhierarchy; memory device 720 being representative of the intermediatetier(L₁) of the memory hierarchy; and memory structure 740 beingrepresentative of the upper tier (L₂) of the memory hierarchy. For thesake of simplicity, only one memory column is shown at each tier, suchthat memory column 702 is intended to be representative of fundamentaltier (L₀), memory column 722 of intermediate tier (L₁), and memorycolumn 742 of upper tier (L₂).

Tier L₀ memory devices, such as memory module 700, are composed ofmultiple memory cells, generally indicated by memory cell 701, which canbe disposed in row, column, or 2-D array (row and column) formats.Memory module 700 is preferred to employ local bit line sensing, localword line decoding, or both, as was described relative to FIGS. 3through 6. In the present example, module MOO includes both local bitline sensing and local word line decoding. Each memory cell M01 in arespective column of memory cells 702 is coupled with local senseamplifier 703 by local bit lines 704 a, 704 b. Although local bit linesensing can be performed on a memory column having a single memory cell,it is preferred that two, or more, memory cells 701 be coupled withlocal sense amplifier 703. Unlike some prior art memory devices whichdispense with local bit line sensing by employing special memory cellswhich provide strong signals at full logic levels, module 700 can use,and indeed is preferred to use, conventional and low-power memory cells701 as constituent memory cells. An advantage of local bit line sensingis that only a limited voltage swing on bit lines 704 a, 704 b may beneeded by local sense amplifier 703 to accurately sense the state ofmemory cell 701, which permits rapid memory state detection andreporting using substantially less power than with prior art designs.

Tier L₀ local sense amplifier 703 detects the memory state of memorycell 701 by coupling the memory state signal to tier L₀ local senseamplifier 703, via local bit lines 704 a, 704 b. It is preferred thatthe memory state signal be a limited swing voltage signal. Amplifier 703transmits a sensed signal representative of the memory state of memorycell 701 to tier L₁ sense amplifier 723 via tier L₀ local senseamplifier outputs 705 a, 705 b, which are coupled with intermediate tierbit lines 724 a, 724 b. It is preferred that the sensed signal be alimited swing voltage signal, as well. In turn, amplifier 723 transmitsa second sensed signal representative of the memory state of memory cell701 to tier L₂ sense amplifier 743, via tier L₁ local sense amplifieroutputs 725 a, 725 b, which are coupled with upper tier bit lines 744 a,744 b. It also is preferred that the second sensed signal be a limitedvoltage swing signal.

Where tier L₂ is the uppermost tier of the memory hierarchy, as isillustrated in the instant example, sense amplifier 743 can be a globalsense amplifier, which propagates a third signal representative ofmemory cell 701 to associated I/O circuitry (not shown)via senseamplifier output lines 746 a, 746 b. Such I/O circuitry can be similarto I/O in FIG. 1. However, the present invention contemplates ahierarchical structure that can consist of two, three, four, or more,tiers of hierarchy. The uppermost tier signal can be a full-swingsignal. In view of FIG. 7, a skilled artisan would realize that “localbit line sensing” occurs at each tier L₀, L₁, and L₂, in the exemplaryhierarchy, and is desirable, for example, because only a limited voltageswing may be needed to report the requested memory state from a lowertier in the hierarchy to the next higher tier.

Hierarchical memory structures also can employ local word line decoding,as illustrated in memory device 740. In FIG. 7, memory device 740 is theuppermost tier (L₂) in the hierarchical memory structure, thus incomingglobal word line signal 746 is received from global word line drivers(not shown) such as global row address decoders 110 in FIG. 1. Incertain preferred embodiments of the present invention, predecoding isemployed to effect rapid access to desired word lines, althoughpredecoding is not required, and may not be desired, at every tier in aparticular implementation. Signal M46 is received by upper tierpredecoder 747, predecoded and supplied to upper tier (L₂) global wordline decoders, such as global word line decoder 748. Decoder M48 iscoupled with local word line decoder 749 by way of upper tier globalword line 750, and selectively activates upper tier local word linedecoder 749. Activated L₂ local decoder M49, in turn, activates L₂ localword line 751, which propagates selected word line signal 726 tointermediate tier (L₁) predecoder 727. Predecoder 727 decodes andactivates the appropriate intermediate tier (L₁) global word linedecoder, such as global word line decoder 728. Decoder 728 is coupledwith, and selectively activates, tier L₁ local word line decoder 729 byway of tier (L₁) global word line 730. Activated L₁ local decoder 729,in turn, propagates a selected word line signal 706 to fundamental tier(L₀) predecoder 707, which decodes and activates the appropriate tier L₀global word line decoder, such as global word line decoder 708.Activated L₀ local decoder 709, in turn, activates L₀ local word line711; and selects memory cell 701 for access. In view of the foregoingdiscussion of hierarchical word line decoding, a skilled artisan wouldrealize that “local word line decoding” occurs at each tier L₀, L₁, andL₂ in the exemplary hierarchy, and is desirable because a substantialreduction in the time and power needed to access selected memory cellscan be realized.

Although local word line decoding within module 700 is shown in thecontext of a single column of memory cells, such as memory columns 702,722, 742, the present invention contemplates that local word linedecoding be performed across two, or more, columns in each of hierarchytiers, with each of the rows in the respective columns employing two ormore local word line decoders, such as local word line decoders 709,729, 749 which are coupled with respective global word line decoders,such as global word line decoders 708, 728, 748 by way of respectiveglobal word lines, such as global word lines 710, 730, 750. However,there is no requirement that equal numbers of rows and columns beemployed at any two tiers of the hierarchical structure. In general,memory device 720 can be composed of multiple memory modules 700, whichfundamental modules 700 can be disposed in row, column, or 2-D array(row and column) array formats. Such fundamental memory modules can besimilar to those illustrated with respect to FIG. 3 through FIG. 6, andcombinations thereof. Likewise, memory device 740 can be composed ofmultiple memory devices 720, which intermediate devices 720 also can bedisposed in row, column, or 2-D array (row and column) formats. Thisextended, and extendable, hierarchality permits the formation ofmultidimensional memory modules that are distinct from prior arthierarchy-like implementations, which generally are 2-D groupings ofbanked, paged, or segmented memory devices, or register file memorydevices, lacking local functionality at each tier in the hierarchy.

Fast Decoder with Asynchronous Reset

Typically, local decoder reset can be used to generate narrow pulsewidths on word lines in a fast memory device. The input signals to theword line decoder are generally synchronized to a clock, or chip select,signal. However, it is desirable that the word line be resetindependently of the clock and also of the varying of the input signalsto the word line decoder.

FIG. 8 is a circuit diagram illustrative of an asynchronously-resettabledecoder 800 according to this aspect of the present invention. It may bedesirable to implement the AND function, for example, by source-coupledlogic. The capacitance on the input x2_n 802 can be generally large,therefore the AND function is performed with about one inverter delayplus three buffer stages. The buffers are skewed, which decreases theload capacitance by about one-half and decreases the buffer delay.

In order to be able to independently reset word line WL 804, it isdesirable that inputs 802, 803 be isolated from output 804, and the node805 should be charged to V_(dd), turning off the large PMOS driver M8807 once word line WL 804 is set to logical HIGH. Charging of node 805to V_(dd) can be accomplished by a feedback-resetting loop. Inputs 802,803 can be isolated from output 804 setting NMOS device 808 to logicLOW. When output WL 804 goes high, monitor node 810 is discharged toground, and device MO 812 is shut-off, thus isolating inputs 802, 803from output WL 804. The feedback loop precharges the rest of the nodesin the buffers via monitor node 810, and PMOSFET M13 815 is turned on,connecting the input x2_n 802 to node 810. Decoder 800 will not fireagain until x2_n 802 is reset to V_(dd), which usually happens when thesystem clock signal changes to logic LOW. Once x2_n 802 is logic HIGH,node 810 charges to V_(dd), with the assistance of PMOS device M14 818,and device MO 812 is turned on. This turns off PMOS device M13 815, thusisolating input x2_n 802 from the reset loop which employs node 810.Decoder 800 is now ready for the next input cycle.

Limited Swing Driver Circuit

FIG. 9 illustrates limited swing driver circuit 900 according to anaspect of the invention herein. In long word length memories, aconsiderable amount of power may be consumed in the data buses. Limitingthe voltage swing in such buses can decrease the overall powerdissipation of the system. This also can be true for a system where asignificant amount of power is dissipated in switching lines with highcapacitance. Limited-swing driver circuit 900 can reduce powerdissipation, for example, in high capacitance lines. When IN signal 902is logic HIGH, NMOS transistor MN1 904 conducts, and node 905 iseffectively pulled to ground. In addition, bitline 910 is dischargedthrough PMOSFET MP1 912. By appropriate device sizing, the voltage swingon bitline 910 can be limited to a desired value, when the inverter,formed by CMOSFETS MP2 914 and MN2 916, switches OFF PMOSFET MP1 912. Ingeneral, the size of circuit 900 is related to the capacitance(C_(bitline)) 918 being driven, and the sizes of MP2 914 and MN2 916. Inanother embodiment of this aspect of the present invention, limitedswing driver circuit includes a tri-state output enable, and aself-resetting feature. Tri-state functionality is desirable when datalines are multiplexed or shared. Although the voltage at memory cellnode 905 can swing to approximately zero volts, it is most desirablethat the bitline voltage swing only by about 200-300 mV.

Single-ended Sense Amplifier with Sample-and-Hold Reference

In general, single-ended sense amplifiers are useful to save metalspace, however, existing designs tend not to be robust due to theirsusceptibility to power supply and ground noise. In yet another aspectof the present invention, FIG. 10 illustrates a single-ended senseamplifier 1000, preferably with a sample-and-hold reference. Amplifier1000 can be useful, for example, as a global sense amplifier, sensinginput data. At the beginning of an operation, DataIn 1004 is sampled,preferably just before the measurement begins. Therefore, supply,ground, or other noise will affect the reference voltage of senseamplifier 1000 generally in the same way noise affects node to bemeasured, tending to increase the noise immunity of the sense amplifier1000. Both inputs 1010, 1011 of differential amplifier 1012 are at thevoltage level of DataIn 1004 when the activate signal (GWSELH) 1014 islogic LOW (i.e., at zero potential). At a preselected interval beforethe measurement begins, but before DataIn 1013 begins to change,activate signal (GWSELH) 1014 is asserted to logic HIGH, therebyisolating the input node 1002 of the transistor M162 1008. The DataInvoltage existing just before the measurement is taken is sampled andheld as a reference, thereby making the circuit substantiallyindependent of ground or supply voltage references. Transistors M1901025 and M187 1026 can add capacitance to the node 1021 where thereference voltage is stored. Transistor M190 1025 also can be used as apump capacitance to compensate for the voltage decrease at the referencenode 1021 when the activate signal becomes HIGH and pulls the source1002 of M162 1008 to a lower voltage. Feedback 1030 from output dataData_toLSA 1035, being transmitted to a local sense amplifier (notshown), is coupled with the source/drain of transistor M187 1026,actively adjusting the reference voltage at node 1021 by capacitivecoupling, thereby adjusting the amplifier gain adaptively.

Sense Ampilifier with Offset Cancellation and Charge-share Limited SwingDrivers

In yet another aspect of the present invention, a latch-type senseamplifier 1100 with dynamic offset cancellation is provided. Senseamplifier 1100 also may be useful as a global sense amplifier, and issuited for use in conjunction with hierarchical bitline sensing.Typically, the sensitivity of differential sense amplifiers can belimited by the offsets caused by inherent process variations for devices(“device matching”), and dynamic offsets that may develop on the inputlines during high-speed operation. Decreasing the amplifier offsetusually results in a corresponding decrease in the minimum bitline swingrequired for reliable operation. Smaller bitline swings can lead tofaster, lower power memory operation. With amplifier 1100, the offset onbitlines can be canceled by the triple PMOS precharge-and-balancetransistors M3 1101, M4 1102, M5 1103, which arrangement is known tothose skilled in the art. However, despite precharge-and-balancetransistors 1101-1103, an additional offset at the inputs of the latchmay exist. By employing balancing PMOS transistor (M14) 1110, any offsetthat may be present at the input of the latch-type differential senseamplifier can be substantially equalized. Sense amplifier 1100demonstrates a charge-sharing limited swing driver 1115. Global bitlines1150, 1151 are disconnected from sense amplifier 1100 when senseamplifier 1100 is not being used, i.e., in a tri-state condition. Senseamplifier 1100 can be in a precharged state if both input/output nodesare logic HIGH, i.e., if both of the PMOS drivers, M38 1130 and M29 1131are off (inputs at logic HIGH). A large capacitor, C₀ 1135, in senseamplifier 1100 can be kept substantially at zero volts by two seriesNMOS transistors, M37 1140 and M40 1141. The size of capacitor 1135 canbe determined by the amount of voltage swing typically needed on globalbitlines 1120, 1121.

When sense amplifier 1100 is activated, and bitlines 1150, 1151 arelogic HIGH, PMOS transistor M29 1131 is turned on and global bit_n 1150is discharged with a limited swing. When a bit to be read is logic LOW,PMOS transistor M38 1130 is turned on, and the global bit 1151 isdischarged with a limited swing. This charge-sharing scheme can resultin very little power consumption, because only the charge that causesthe limited voltage swing on the global bitlines 1150, 1151 isdischarged to ground. That is, there is substantially no “crowbar”current. Furthermore, this aspect of the present invention can be usefulin memories where the global bitlines are multiplexed for input andoutput.

Module-tier Memory Redundancy Implementation

In FIG. 12, memory structure 1200, composed of hierarchical functionalmemory modules 1201 is preferred to have at least one or more redundantmemory rows 1202, 1204; one, or more redundant memory columns 1206,1208; or both, within each module 1201. It is preferred that theredundant memory rows 1202, 1204, and/or columns 1206, 1208 be paired,because it has been observed that bit cell failures tend to occur inpairs. Module-level redundancy, as shown in FIG. 12, where redundancy isimplemented using a preselected number of redundant memory rows 1202,1204, or redundant memory columns 1206, 1208, within memory module 1201,can be a very area-efficient approach provided the typical number of bitcell failures per module remains small. By implementing only a singlerow 1202 or a single column 1206 or both in memory module 1201, only oneadditional multiplexer is needed for the respective row or column.Although it may be simpler to provide redundant memory cell circuitsthat can be activated during product testing during the manufacturingstage, it may also be desirable to activate selected redundant memorycells when the memory product is in service, e.g., during maintenance oron-the-fly during product operation. Such activation can be effected bynumerous techniques and support circuitry which are well-known in theart.

Redundant Module Memory Redundancy Implementation

As shown in FIG. 13, memory redundancy also may be implemented byproviding redundant module 1301 to memory structure 1300, which iscomposed of primary modules 1304, 1305, 1306, 1307. Redundant module1301 can be a one-for-one replacement of a failed primary module, e.g,module 1304. In another aspect of the invention, redundant module 1301may be partitioned into smaller redundant memory segments 1310 a-d withrespective ones of segments 1310 a-d being available as redundant memorycells, for example, for respective portions of primary memory modules1304-1307 which have failed. The number of memory cells assigned to eachsegment 1310 a-d in redundant memory module 1301, may be a fixed number,or may be flexibly allocatable to accommodate different numbers offailed memory circuits in respective primary memory modules 1304-1307.

Memory Redundancy Device

FIG. 14 illustrates another aspect of the present invention whichprovides an implementation of row and column redundancy for a memorystructure such as memory structure 100 in FIG. 1, or memory structure300 in FIG. 3. This aspect of the present invention can be implementedby employing fuses that are programmable, for example, duringproduction. Examples of such uses include metal fuses that are blownelectrically, or by a focused laser; or a double-gated device, which canbe permanently programmed. Although the technique can be applied toprovide row redundancy, or column redundancy, or both, the presentdiscussion will describe column redundancy in which both inputs andoutputs may need the advantages of redundancy.

FIG. 14 shows an embodiment of this aspect of the invention hereinhaving four pairs of columns 1402 a-d with one redundant pair 1404. Itis desirable to implement this aspect of the present invention as pairsof lines because a significant number of RAM failures occur in pairs,whether column or row. Nevertheless, this aspect of the presentinvention also contemplates single line redundancy. In general, thenumber of fuses in fuse box 1403 used to provide redundancy can belogarithmically related to the number line pairs, e.g., column pairs:log₂ (number of column pairs), where the number of column pairs includesthe redundant pairs as well. Because fuses tend to be large, theirnumber should be minimized, thus the logarithmic relation isadvantageous. Fuse outputs 1405 are fed into decoder circuits 1406 a-d,e.g., one fuse output per column pair. A fuse output creates what isreferred to herein as a “shift pointer”. The shift pointer indicates theshift signal in the column pair to be made redundant, and subsequentcolumn pairs can then be inactivated. It is desirable that the signals1405 from fuse box 1410 are decoded to generate shift signal 1412 a-d ateach column pair. When shift signal 1412 a-d for a particular columnpair 1402 a-d location is selected, as decoded from fuse signals 1405,shift pointer 1412 a-d is said to be pointing at this location. Theshift signals for this column, and all subsequent columns to the rightof the column of pair shift pointer also become inactive.

This aspect of the present invention can be illustrated additionally inFIG. 15A and FIG. 15B, by way of the aforementioned concept of “shiftpointers.” In FIG. 15A, three column pairs 1501, 1502, 1503, and oneredundant column pair 1504 are shown. The shift procedure isconceptually indicated by way of “line diagrams”. The top lines1505-1508 of the line diagrams are representative of columns 1501-1504within the memory core while bottom line pairs 1509-1511 are the datainput/output pairs from the input/output buffers. When a shift signal,such as a signal 1405 in FIG. 14, for a particular column pair 1501-1503is logical LOW, it is preferred that the data in 1509-1511 be connectedto respective column 1501-1503 directly above it by multiplexers. FIG.15B is illustrative of having a failed column state. When shift signalis logical HIGH, such as a signal 1405 in FIG. 14, a failed column isindicated, such as column 1552. Active columns 1550, 1551 remainunfaulted, and continue to receive their data via I/O lines 1554, 1555.However, because column 1552 has failed, data from I/O buffer 1556 canbe multiplexed to the redundant column pair 1553. Diagrammatically, itappears that data in are shifted left while data out from the memorycore columns are shifted right. By adjusting the location of the shiftpointer, which generally is determined by the state of the fuses, theunused redundant column pair can be shifted to coincide with anonfunctional column, e.g., column 1552, thereby repairing the columnfault and boosting the fully functional memory yield.

Selector for Redundant Memory Circuits

FIG. 16 illustrates yet another aspect of the present invention, inwhich selector 1600 is adapted to provide a form of redundancy. Selector1600 can include a primary decoder circuit 1605, which may be a globalword line decoder, which is coupled with a multiplexer 1610. MUX 1610can be activated by a redundancy circuit 1620, which may be a fusesystem, programable memory, or other circuit capable of providing anactivation signal 1630 to selector 1600 via MUX 1610. Selector 1600 issuitable for implementing module-level redundancy, such as thatdescribed relative to module 1200 in FIG. 12, which may be rowredundancy or column redundancy for a given implementation. In theordinary course of operation, input word line signal 1650 is decoded indecoder circuit 1605 and, in the absence of a fault on local word line1670, the word line signal is passed to first local line 1680. In theevent a fault is detected, MUX 1610, selects second local line 1660,which is preferred to be a redundant word line.

Fast Decoder with Row Redundancy

FIG. 17 illustrates a preferred embodiment of selector 1600 in FIG. 16,in the form of decoder 1700 with row redundancy as realized in ahierarchical memory environment. Decoder 1700 may be particularlysuitable for implementing module-level redundancy, such as thatdescribed relative to module 1200 in FIG. 12. Global decoder 1700, canoperate similarly to the manner of asynchronously-resettable decoder 800of FIG. 8. In general, decoder 1700 can be coupled with a first,designated memory row, and a second, alternative memory row. Althoughthe second row may be a physical row adjacent the first memory row, andanother of the originally designated rows of the memory module, thesecond row also may be a redundant row which is implemented in themodule. Although row decoder 1700 decodes the first memory row undernormal operations, it also is disposed to select and decode the secondmemory row in responsive to an alternative-row-select signal. Where thesecond row is a redundant row, it may be more suitable to deem theselection signal to be a “redundant-row-select” signal. Theaforementioned row select signals are illustrated as inputs 1701 and1702.

Thus, when input 1701 or 1702 is activated, decoder 1700 transfers thelocal word line signal, usually output on WL 1706, to be output onxL_Next 1705, which is coupled with an adjacent word line. In general,when a word line decoder, positioned at a particular location in amemory module, receives a shift signal, the remaining decoderssubsequent to that decoder also shift, so that the last decoder in thesequence shifts its respective WL data to a redundant word line. Using atwo-dimensional conceptual model where a redundant row is at the bottomof a model, this process may be described as having a fault at aparticular position effect a downward shift of all local word lines atand below the position of the fault. Those local word lines above theposition of the fault can remain unchanged.

Hybrid Single Port and Dual Port (R/W) Functionality

Hierarchical memory module implementations realize significant timesavings due in part to localized functionality. Signal propagation timesat the local module tier tend to be substantially less than the typicalaccess time of a larger memory structure, even those employing existingpaged, banked, and segmented memory array, and register file schemes.Indeed, both read and write operations performed at the fundamentalmodule tier can occur within a fraction of the overall memory structureaccess time. Furthermore, because bitline sensing, in accordance withthe present invention, is power-conservative, and does not result in asubstantial decay of precharge voltages, the bitline voltage levelsafter an operation tend to be marginally reduced. As a result, incertain preferred embodiments of the present invention, it is possibleto perform two operations back-to-back without an intervening pre-chargecycle, and to do so within a single access cycle of the overall memorystructure. Therefore, although a memory device may be designed as to besingle-port device, a preferred memory module embodiment functionssimilarly to a two-port memory device, which can afford such anembodiment a considerable advantage over prior art memory structures ofcomparable overall memory size.

FIG. 18 illustrates one particular embodiment of this aspect of thepresent invention, in memory structure 1800, where both local bitlinesensing and local word line decoding are used, as described above.Memory structure 1800 includes memory module 1805 which is coupled withlocal word line decoder 1815 and local bit sense amplifier 1820. Withinmemory module 1805 are a predefined number of memory cells, for example,memory cell 1825, which is coupled with local word line decoder 1815 vialocal word line 1810, and local bit sense amplifier 1820 via localbitlines 1830. With typical single-port functionality, local bitlines1830 are precharged prior to both READ and WRITE operations. During atypical READ operation, predecoder 1835 activates the appropriate globalword line decoder 1840, which, in turn, activates local word linedecoder 1815. Once local word line decoder 1815 determines thatassociated memory cell 1825 is to be evaluated, it opens memory cell1825 for evaluation, and activates local bit sense amplifier 1820. Atthe end of the local sensing period, local bit sense amplifier 1820outputs the sensed data value onto global bitlines 1845. After globalsense amplifier 1850 senses the data value, the data is output to theI/O buffer 1855. If a WRITE operation is to follow the READ operation, atypical single-port device would perform another precharge operationbefore the WRITE operation can commence.

In this particular embodiment of dual-port functionality, thepredecoding step of a subsequent WRITE operation can commenceessentially immediately after local bitline sense amplifier 1820completes the evaluation of memory cell 1825, that is, at the inceptionof sensing cycle for global sense amplifier 1850, and prior to the databeing available to I/O buffer 1855. Thus, during the period encompassingthe operation of global sense amplifier 1850 and I/O buffer 1855, andwhile the READ operation is still in progress, predecoder 1835 canreceive and decode the address signals for a subsequent WRITE operation,and activate global word line decoder 1840 accordingly. In turn, globalword line decoder 1840 activates local word line 1815 in anticipation ofthe impending WRITE operation. As soon as the datum is read out of I/Obuffer 1855, the new datum associated with the WRITE cycle can beadmitted to I/O buffer 1855 and immediately written to, for example,memory cell 1825, without a prior precharge cycle. In order to providethe memory addresses for these READ and WRITE operations in a mannerconsistent with this embodiment of the invention, it is preferred thatthe clocking cycle of predecoder 1810 be faster than the access cycle ofthe overall memory structure 1800. For example, it may be desirable toadapt the predecoding clock cycle to be about twice, or perhaps greaterthan twice, the nominal access cycle for structure 1800. In this manner,a PRECHARGE-READ-WRITE operation can be performed upon the same memorycell within the same memory module in less than one access cycle,thereby obtaining dual-port functionality from a single port device. Italso is contemplated that the aforementioned embodiment can be adaptedto realize three or more operations within a single access cycle, aspermitted by the unused time during an access cycle.

Fortuitously, the enhanced functionality described above is particularlysuited to large memory structures with comparatively small constituentmodules, where the disparity between global and local access times ismore pronounced. Moreover, in environments where delays due to signalpropagation across interconnections, and to signal propagation delaysthrough co-embedded logic components may result in sufficient idle timefor a memory structure, this enhanced functionality may advantageouslymake use of otherwise “wasted” time.

FIG. 19 illustrates high precision delay measurement (HPDM) circuit1900, according to one aspect of the present invention, which canprovide timing measurements of less than that of a single gate delay,relative to the underlying technology. These measurements can be, forexample, of signal delays and periods, pulse widths, clock skews, etc.HPDM circuit 1900 also can provide pulse, trigger, and timing signals toother circuits, including sense amplifiers, word line decoders, clockdevices, synchronizers, state machines, and the like. Indeed, HPDMcircuit 1900 is a measurement circuit of widespread applicability. Forexample, HPDM circuit 1900 can be implemented within a high-performancemicroprocessor, where accurate measurement of internal time intervals,perhaps on the order of a few picoseconds, can be very difficult usingdevices external to the microprocessor. HPDM circuit 1900 can be used toprecisely measure skew between and among signals, and thus also can beused to introduce or eliminate measured skew intervals. HDPM circuit1900 also can be employed to characterize the signals of individualcomponents, which may be unmatched, or poorly-matched components, aswell as to bring such components into substantial synchrony.Furthermore, HPDM circuit 1900 can advantageously be used in registerfiles, transceivers, adaptive circuits, and a myriad of otherapplications in which precise interval measurement is desirable initself, and in the context of adapting the behavior of components,circuits, and systems, responsive to those measured intervals.Advantageously, HPDM circuit 1900 can be devised to be responsive tooperating voltage, design and process variations, design rule scaling,etc., relative to the underlying technology, including, withoutlimitation, bipolar, nMOS, CMOS, BiCMOS, and GaAs technologies. Thus, anHPDM circuit 1900 designed to accurately measure intervals relevant to1.8 micron technology will scales in operation to accurately measureintervals relevant to 0.18 micron technology. Although HPDM circuit 1900can be adapted to measure fixed time intervals, and thus remainindependent of process variations, design rule scaling, etc., it ispreferred that HPDM circuit 1900 be allowed to respond to the technologyand design rules at hand. In general, the core of an effective HPDMcircuit capable of measuring intervals on the order of picoseconds, canrequire only a few scores of transistors which occupy a minimalfootprint. This is in stark contrast to its counterpart in thehuman-scale domain, i.e., a an expensive, high-precision handheld, orbench side, electronic test device.

One feature of HPDM circuit 1900 is modified ring oscillator 1905. As iswell-known in the art of ring oscillators, the oscillation period, T₀,of a ring oscillator having N stages is approximately equal to 2NT_(D),where T_(D) is the large-signal delay of the gate/inverter of eachstage. The predetermined oscillation period, T₀, can be chosen byselecting the number of gates to be employed in the ring oscillator. Ingeneral, T_(D) is a function of the rise and fall times associated witha gate which, in turn, are related to the underlying parametersincluding, for example, gate transistor geometries and fabricationprocess. These parameters are manipulable such that T_(D) can be tunedto deliver a predetermined gate delay time. In a preferred embodiment ofthe present invention in the context of a specific embodiment of ahierarchical memory structure, it is desirable that the parameters berelated to a CMOS device implementation using 0.18 micron (μm) designrules. However, a skilled artisan would realize that HPDM circuit 1900is not limited thereto, and can be employed in other technologies,including, without limitation, bipolar, nMOS, CMOS, BiCMOS, GaAs, andSiGe technologies, regardless of design rule, and irrespective ofwhether implemented on Si substrate, SOI and its variants, etc.

Although exemplary HPDM circuit 1900 employs seven (7) stage ringoscillator 1905, a greater or lesser number of stages may be used,depending upon the desired oscillation frequency. In this example, ringoscillator 1905 includes NAND gate 1910, the output of which beingdesignated as the first stage output 1920; and six inverter gates,1911-1916, whose outputs 1921-1926 are respectively designated as thesecond through seventh stage outputs.

In addition to ring oscillator 1905, HPDM circuit 1900 can includememory elements 1930-1937, each of which being coupled with apreselected oscillator stage. The selection and arrangement of memoryelements 1930-1937, make it possible to measure a minimum time quantum,T_(L), which is accurate to about one-half of a gate delay, that is,T_(L)≈T_(D)/2. The maximum length of time, T_(M), that can usefully bemeasured by HPDM circuit 1900 is determinable by selecting one or morememory devices, or counters, to keep track of the number of oscillationcycles completed since the activation of oscillator 1905, for example,by ENABLE signal 1940. Where the selected counter is a single 3-bitdevice, for example, up to eight (8) complete cycles through oscillator1905 can be detected, with each cycle being completed in T₀ time.Therefore, using the single three-bit counter as an example,T_(M)≈8T_(o). The remaining memory elements 1932-1937 can be used toindicate the point during a particular oscillator cycle at which ENABLEsignal 1940 was deactivated, as determined by examining the respectivestates of given memory elements 1932-1937 after deactivation ofoscillator 1905.

In HPDM circuit 1900, it is preferred that a k-bit positiveedge-triggered counter (PET) 1930, and a k-bit negative edge-triggeredcounter (NET) 1931, be coupled with first stage output 1920. Further, itis preferred that a dual edge-triggered counter (DET) 1932-1937 becoupled with respective outputs 1921-1925 of Oscillator 1905. In aparticular embodiment of the invention, PET 1930 and NET 1931 are eachselected to be three-bit counters (i.e., k=3), and each of DET 1932-1937are selected to be one-bit counters (latches). An advantage of usingdual edge detection in counters 1932-1937 is that the edge of aparticular oscillation signal propagating through ring oscillator 1905can be registered at all stages, and the location of the oscillationsignal at a specific time can be determined therefrom. Because apropagating oscillation signal alternates polarity during sequentiallysubsequent passages through ring oscillator 1905, it is preferred toemploy both NET circuit 1930 and PET 1931, and that the negative edge ofa particular oscillation signal be sensed as the completion of the firstlooping event, or cycle, through ring oscillator 1905.

The operation of HPDM circuit 1900 can be summarized as follows: withEnableL signal 1904 asserted HIGH, ring oscillator 1905 is in the STATICmode, so that setting ResetL signal 1906 to LOW resets counters1930-1937. By setting StartH signal 1907 to HIGH, sets RS flip-flop 1908which, in turn, sets ring oscillator 1905 to the ACTIVE mode bypropagating an oscillation signal. Each edge of the oscillation signalcan be traced by identifying the switching activity at each stage output1920-1926. PET 1930 and NET 1931, which sense first stage output 1920identify and count looping events. It is preferred that the maximumdelay to be measured can be represented by the maximum count of PET 1930and NET 1931, so that the counters do not overflow. To stop thepropagation of the oscillation signal through ring oscillator 1905,StopL signal 1909 is set LOW, RS flip-flop 1908 is reset, and ringoscillator 1905 is returned to the STATIC mode of operation. Also, thedata in counters 1930-1937 are isolated from output stages 1920-1926 bysetting enL signal 1950 to LOW and enH signal 1951 to HIGH. The digitaldata is then read out through ports lpos 1955, lneg 1956, and del 1957.With knowledge of the average stage delay, the digital data then can beinterpreted to provide an accurate measurement, in real time units, ofthe interval during which ring oscillator 1905 was in the ACTIVE mode ofoperation. HPDM circuit 1900 can be configured to provide, for example,a precise clock or triggering signal, such as TRIG signal 1945, afterthe passage of a predetermined quantum of time. Within the context of amemory system, such quantum of time can be, for example, the timenecessary to sense the state of a memory cell, to keep active awordline, etc.

The average stage delay through stages 1910-1916 can be determined byoperating ring oscillator 1905 for a predetermined averaging time byasserting StartH 1907 and StopL 1909 to HIGH, thereby incrementingcounters 1930-1937. In a preferred embodiment of the present invention,the overflow of NET 1931 is tracked, with each overflow event beingindicative of 2^(k) looping events through ring oscillator 1905. It ispreferred that this tracking be effected by a divider circuit, forexample, DIVIDE-BY-64 circuit 1953. At the end of the predeterminedaveraging time, data from divider 1953 may be read out through portRO_div64 1954 as a waveform, and then analyzed to determine the averageoscillator stage delay. However, a skilled artisan would realize thatthe central functionality of HPDM circuit 1900, i.e., to provide precisemeasurement of a predetermined time quantum, would remain unaltered ifDIVIDE-BY-64 circuit 1953, or similar divider circuit, were not includedtherein.

HPDM circuit 1900 can be used for many timing applications whether ornot in the context of a memory structure, for example, to preciselyshape pulsed waveforms and duty cycles; to skew, de-skew across one ormore clocked circuits, or to measure the skew of such circuits; toprovide high-precision test data; to indicate the beginning, end, orduration of a signal or event; and so forth. Furthermore, HPDM circuit1900 can be applied to innumerable electronic devices other than memorystructures, where precise timing measurement is desired.

Accurate self-timed circuits are important features of robust, low-powermemories. Replica bitline techniques have been described in the priorart to match the timing of control circuits and sense amplifiers to thememory cell characteristics, over wide variations in process,temperature, and operation voltage. One of the problems with some priorart schemes is that split dummy bitlines cluster word-lines togetherinto groups, and thus only one word-line can be activated during amemory cycle. Before a subsequent activation of a word-line within thesame group, the dummy bitlines must be precharged, creating anundesirable delay. The diffusion replica delay technique of the presentinvention substantially matches the capacitance of a dummy bitline byusing a diffusion capacitor, preferably for each row. Some prior arttechniques employed replica bit-columns which can add to undesirableoperational delays. FIG. 20 illustrates the diffusion replica timingcircuit 2000 which includes transistor 2005 and diffusion capacitance2010. It is desirable that transistor 2005 be an NMOSFET transistorwhich, preferably, is substantially identical to an access transistorchain, if such is used in the memory cells of the memory structure (notshown). It also is desirable that the capacitance of diffusion capacitor2010 is substantially matched to the capacitance of the associatedbitline (not shown). This capacitance can be a predetermined ratio ofthe total bitline capacitance, with the ratio of the diffusioncapacitance to total bitline capacitance remaining substantiallyconstant over process, temperature and voltage variations. The totalbitline capacitance can include both the bitline metal and diffusioncapacitances. In this fashion, all rows in a memory device which usetiming circuit 2000 can be independently accessible with substantiallyfully-operation self-timing, even when another row in the same memorymodule has been activated, and is not yet precharged. Thus,write-after-read operations may be multiplexed into a memory modulewithout substantial access time or area penalties. Thus, it is desirableto employ diffusion replica delay circuit 2000 in a memory structuresuch as memory structure 1800, described in FIG. 18. Diffusion replicadelay circuit 2000 can be used to determine the decay time of a bitlinebefore a sense amplifier is activated, halting the decay on the bitline.In this manner, bitline decay voltage can be limited to a relativelysmall magnitude, thus saving power and decreasing memory access time.Furthermore, timing circuit 2000 can be used to accurately generate manytiming signals in a memory structure such as structure 1800 in FIG. 18,including, without limitation, precharge, write, and shut-off timingsignals.

FIG. 21 illustrates an embodiment of the diffusion replica delay circuit2000 in FIG. 20. Word-line activation of a memory cell frequency ispulsed to limit the voltage swing on the high capacitance bitlines, inorder to minimize power consumption, particularly in wide word lengthmemory structures. In order to accurately control the magnitude of abitline voltage swing, dummy bitlines can be used. It is desirable thatthese dummy bitlines have a capacitance which is a predefined fractionof the actual bitline capacitance. In such a device, the capacitanceratio between dummy bitlines and real bitlines can affect the voltageswing on the real bitlines. In prior art devices using dummy bitlines, aglobal dummy bitline for a memory block having a global reset loop hasbeen utilized. Such prior art schemes using global resetting tends todeliver pulse widths of a duration substantially equivalent to the delayof global word-line drivers. Such an extend pulse width allows for abitline voltage swing which can be in excess of what actually isrequired to activate a sense amplifier. This is undesirable in fastmemory structures, because the additional, and unnecessary, voltageswing translates into a slower structure with greater powerrequirements. In one aspect of the present invention, dummy bitlines arepreferably partitioned such that the local bitlines generally exhibit asmall capacitance and a short discharge time. Word-line pulse signals ofvery short duration (e.g., 500 ps or less) are desirable in order tolimit the bitline voltage swing. It also may be desirable to providelocal reset of split dummy bitlines to provide very short word-linepulses. Replica word-line 2110 can be used to minimize the delay betweenactivation of memory cell 2120 and related sense amplifier 2130. Suchlocal signaling is preferred over global signal distribution onrelatively long, highly capacitive word-lines. Word-line 2140 activatesdummy cell 2150 along with associated memory cell 2120, which is to beaccessed. Dummy cell 2150 can be part of dummy column 2160 which may besplit into small groups (for example, eight or sixteen groups). The sizeof each split dummy group can be changed to adjust the voltage swing onthe bitline. When a dummy bitline is completely discharged, reset signal2170 can be locally generated which pulls word-line 2140 substantiallyto ground.

FIG. 22A illustrates controlled voltage swing data bus circuit (CVS)2200 which can be useful in realizing lower power, high speed, and denseinterconnection buses. CVS 2200 can reduce bus power consumption byimposing a limited, controlled voltage swing on bus 2215. In anessential configuration, CVS 2000 can include inverter 2205, pMOS passtransistor T2 2210, and one nMOS discharge transistor, such astransistor T1 a 2205 a. Both transistors T1 a 2205 a, and T2 2210 can beprogrammed to control the rate and extent of voltage swings on bus 2215such that a first preselected bus operational characteristic is providedin response to input signal 2220 a. Additional discharge transistors T1b 2205 b and T1 c 2205 c can be coupled with pass transistor T2 2210,and individually programmed to respectively provide a second preselectedbus operational characteristic, as well as a third preselected busoperational characteristic, responsive to respective input signals 2220b, 2220 c. The preselected bus operational characteristic can be forexample, the rate of discharge of the bus voltage through the respectivedischarge transistor T1 a 2205 a, T1 b 2205 b, and T1 c 2205 c, suchthat bus 2215 is disposed to provide encoded signals, or multilevellogic, thereon. For example, as depicted in FIG. 22A, CVS 2200 canprovide three distinct logic levels. Additional discharge transistors,programmed to provide yet additional logic levels also may be used.Thus, it is possible for bus 2215 to replace two or more lines.Concurrently with effecting a reduction in power consumption, thelimited bus voltage swing advantageously tends to increase the speed ofthe bus.

FIG. 22B illustrates a bidirectional data bus transfer circuit (DBDT)2250 which employs cross-linked inverters I1 2260 and I2 2270 to coupleBUS 1 2252 with BUS 2 2254. It is desirable to incorporate a clockedcharge/discharge circuit with DBDT 2250. Coupled with inverter I1 2260is clocked charge transistor MPC1 2266 and clocked discharge transistorMNC1 2268. Similarly, inverter I2 2270 is coupled with clocked chargetransistor MPC2 2276 and clocked discharge transistor MNC2 2278.Transistors MPC1 2266, MNC1 2268, MPC2 2276, and MNC2 2278 are preferredto be driven by clock signal 2280.

Beginning with clock signal 2280 going LOW, charge transistors MPC1 2266and MPC2 2276 turn ON, allowing BUS 1 input node 2256 and BUS 2 inputnode 2258 to be precharged to HIGH. Additionally, discharge transistorsMNC1 2268 and MNC2 2278 are turned OFF, so that no substantial dischargeoccurs. By taking input nodes 2256, 2258 to HIGH, respective signalspropagate through, and are inverted by inverters I1 2260 and I2 2270providing a LOW signal to BUS 1 pass transistor MP12 2262 and BUS 2 passMP22 2272, respectively, allowing the signal on BUS 1 2252 to beadmitted to input node 2256, and then to pass through to BUS2 input node2258 to BUS 2 2254, and vice versa. When clock signal 2280 rises toHIGH, both charge transistors MPC1 2266 and MPC2 2276 turn OFF, anddischarge transistors MNC1 2268 and MNC2 2278 turn ON, latching the dataonto BUS 1 2252 and BUS 2 2254. Upon the next LOW phase of clock signal2280, a changed signal value on either BUS 1 2252 or BUS 2 2254 willpropagate between the buses.

Many alterations and modifications may be made by those having ordinaryskill in the art without departing from the spirit and scope of theinvention. Therefore, it must be understood that the illustratedembodiments have been set forth only for the purposes of example, andthat it should not be taken as limiting the invention as defined by thefollowing claims. The following claims are, therefore, to be read toinclude not only the combination of elements which are literally setforth but all equivalent elements for performing substantially the samefunction in substantially the same way to obtain substantially the sameresult. The claims are thus to be understood to include what isspecifically illustrated and described above, what is conceptuallyequivalent, and also what incorporates the essential idea of theinvention.

What is claimed is:
 1. A controlled voltage swing data bus circuit fortransferring data to a data bus, comprising: a. a pass transistorcoupled with the bus; b. a discharge transistor coupled with the passtransistor and ground, the discharge transistor programmed to impose afirst preselected bus operational characteristic on the data bus; and c.an inverter coupled between the discharge transistor and the passtransistor, the inverter and the discharge transistor forming a signalnode, the inverter selectably driving the pass transistor gate,responsive to a signal node voltage value.
 2. The controlled voltageswing data bus circuit of claim 1, wherein the pass transistor isprogrammed to impose a second preselected bus operational characteristicon the data bus.
 3. The controlled voltage swing data bus circuit ofclaim 1, further comprising a plurality of discharge transistors, onesof the plurality of discharge transistors being selectively programmedto impose respective preselected bus operational characteristics on thedata bus.
 4. The controlled voltage swing data bus circuit of claim 1,wherein the pass transistor couples the data bus with one of a globalsense amplifier, a local sense amplifier, a global wordline decoder, anda local wordline decoder.
 5. The controlled voltage swing data buscircuit of claim 2, wherein the pass transistor couples the data buswith one of a global sense amplifier, a local sense amplifier, a globalwordline decoder, and a local wordline decoder.
 6. A controlled voltageswing data bus transfer circuit for transferring data between a firstdata bus and a second data bus, comprising: a. a first inverter; b. afirst pass transistor, coupled between the first data bus and the firstinverter, the coupling of the first pass transistor and the firstinverter forming a first signal node; c. a second inverter cross-linkedto the first inverter; d. a second pass transistor, coupled between thesecond data bus and the second inverter, the coupling of the second passtransistor and the second inverter forming a second signal node; e. afirst signal discharge transistor coupled between the first signal nodeand ground; and f. a second signal discharge transistor coupled betweenthe second signal node and ground, wherein the first inverter transfersa signal representative of first data on the first data bus to thesecond signal discharge transistor and the second inverter transfers asignal representative of second data on the second data bus to the firstsignal discharge transistor.
 7. The controlled voltage swing data bustransfer circuit of claim 6, further comprising: a. a first node chargetransistor, coupled between V_(dd) and the first signal node; b. a firstnode discharge transistor coupled between the first signal dischargetransistor and ground; c. a second charge transistor, coupled betweenV_(dd) and the second signal node; d. a second node discharge transistorcoupled between the second signal discharge transistor and ground,wherein each of the first and second node charge transistors pull therespective first and second signal node to V_(dd), when a LOW gatesignal is applied thereto.
 8. The controlled voltage swing data bustransfer circuit of claim 7, further comprising a clocking signaloperably coupled with the first node charge transistor, the first nodedischarge transistor, the second node charge transistor, and the secondnode discharge transistor.
 9. The controlled voltage swing data bustransfer circuit of claim 6, wherein one of the first data bus and thesecond data bus is coupled with one of a global sense amplifier, a localsense amplifier, a global wordline decoder, and a local wordlinedecoder.
 10. The controlled voltage swing data bus transfer circuit ofclaim 8, wherein one of the first data bus and the second data bus iscoupled with one of a global sense amplifier, a local sense amplifier, aglobal wordline decoder, and a local wordline decoder.